Magnetic component, power converter and power supply system

ABSTRACT

A magnetic component has a core on which windings are wound. The windings are electrically connected in series to constitute a coil of a first reactor. The winding constitutes a coil of a second reactor. The core has a leg portion on which the winding is wound, a leg portion on which the winding is wound, and a leg portion on which the winding is wound. When a current flows through the windings, magnetic fluxes produced from the windings, respectively, and flowing through the winding counteract each other. Furthermore, when a current flows through the winding, induced voltages produced from the windings, respectively, by the magnetic flux produced by the winding counteract each other.

TECHNICAL FIELD

The present invention relates to a magnetic component, a power converterand a power supply system, and more particularly to a magnetic componentin which two reactors included in different current paths are integratedas well as a power converter and a power supply system including themagnetic component.

BACKGROUND ART

An inductive element used for a power converter is generally formed bywinding coils on a core made of a magnetic material, which is likely toincrease in size. It has therefore been proposed to form a plurality ofreactors integrally by a single magnetic component in a circuitincluding a plurality of inductive elements.

For example, Japanese Patent Laying-Open No. 2009-59995 (PTL 1)describes a configuration of a composite magnetic component in which atransformer and a reactor are integrated. In the configuration describedin PTL 1, a primary winding and a secondary winding are wound on one oftwo cores constituting the transformer, and an extension of the primarywinding is wound on the other core. Accordingly, the reactor connectedin series with the primary winding of the transformer can be formedintegrally with the transformer by means of the above-mentionedextension.

Japanese Patent Laying-Open No. 2009-284647 (PTL 2) describes aconfiguration of a composite transformer in which first and secondinductors and a transformer are formed integrally.

CITATION LIST Patent Literature

PTL 1: Japanese Patent Laying-Open No. 2009-59995

PTL 2: Japanese Patent Laying-Open No. 2009-284647

SUMMARY OF INVENTION Technical Problem

As a mode of power converter, there exists a circuit configurationhaving two reactors respectively included in current paths independentlycontrolled in current. In such a circuit, if an induced voltage isproduced in one of the reactors by the current flowing through the otherreactor, each current can no longer be controlled independently.Therefore, it is a subject for the configuration in which these reactorsare integrated to consider preventing an induced voltage from beingproduced in one of the reactors by the current flowing through the otherreactor.

In the composite magnetic component of PTL 1, the transformer andreactor connected in series are integrated. That is, magnetic elementsincluded in a common current path are integrated, which does not presenta solution for the above-described subject.

In the composite transformer of PTL 2, the current paths in the firstand second reactors are included on the primary side and secondary sideof the transformer. Therefore, the composite transformer of PTL 2 isapplied to a circuit aiming at causing induced voltages to act on eachother between the current path in the first reactor and the current pathin the second reactor. Hence, PTL 2 also fails to disclose an integratedstructure that can solve the above-described subject.

The present invention was made to solve such a problem, and has anobject to integrally form two reactors respectively included in currentpaths independently controlled in current, thereby achieving reductionin size and weight of a magnetic component as well as a power converterand a power supply system including the magnetic component.

Solution to Problem

In an aspect of the present invention, a magnetic component includesfirst and second windings electrically connected in series through whicha first current flows, a third winding through which a second currentflows, and a core. The core is configured to include a first section onwhich the first winding is wound, a second section on which the secondwinding is wound, and a third section on which the third winding iswound. The first to third windings are wound on the first to thirdsections, respectively, such that, in a state where magnetic saturationdoes not occur in said core, (i) a magnetic flux produced from the firstwinding and flowing through the third winding and a magnetic fluxproduced from the second winding and flowing through the third windingcounteract each other, when the first current flows through the firstand second windings, and (ii) induced voltages produced in the first andsecond windings, respectively, by a magnetic flux produced from thethird winding counteract each other, when the second current flowsthrough the third winding.

In another aspect of the present invention, a magnetic componentincludes first and second windings electrically connected in seriesthrough which a first current flows, a third winding through which asecond current flows, and a core. The core is configured to include afirst section on which the first winding is wound, a second section onwhich the second winding is wound, and a third section on which thethird winding is wound. The first and second windings are wound on thefirst and second sections, respectively, such that the first and secondwindings have winding directions opposite to each other. The core onwhich the first to third windings are wound is configured such that, ina state where magnetic saturation does not occur in the core, a magneticresistance of a first magnetic circuit passing through the first sectionand a magnetic resistance of a second magnetic circuit passing throughthe second section are equivalent, when the second current flows throughthe third winding.

Preferably, in the magnetic component, in the core on which the first tothird windings are wound, in a state where magnetic saturation does notoccur in the core, (i) a magnetic flux produced from the first windingand flowing through the third winding and a magnetic flux produced fromthe second winding and flowing through the third winding counteract eachother, when the first current flows through the first and secondwindings, and (ii) induced voltages produced in the first and secondwindings, respectively, by a magnetic flux produced from the thirdwinding counteract each other, when the second current flows through thethird winding.

In still another aspect of the present invention, a power converterincludes first and second reactors electrically connected between a DCpower source and a load, and a plurality of switching elements arrangedsuch that a first current flowing through the first reactor and a secondcurrent flowing through the second reactor are controlled independently.The first and second reactors are formed integrally by a single magneticcomponent. The magnetic component includes first and second windingselectrically connected in series through which a first current flows, athird winding through which the second current flows, and a core. Thecore is configured to include a first section on which the first windingis wound, a second section on which the second winding is wound, and athird section on which the third winding is wound. The first to thirdwindings are wound on the first to third sections, respectively, suchthat, in a state where magnetic saturation does not occur in said core,(i) a magnetic flux produced from the first winding and flowing throughthe third winding and a magnetic flux produced from the second windingand flowing through the third winding counteract each other, when thefirst current flows through the first and second windings, and (ii)induced voltages produced in the first and second windings,respectively, by a magnetic flux produced from the third windingcounteract each other, when the second current flows through the thirdwinding.

Preferably, in the power converter, in the core on which the first tothird windings are wound, in a state where magnetic saturation does notoccur in said core, when the second current flows through the thirdwinding, a magnetic resistance of a first magnetic circuit passingthrough the first section and a magnetic resistance of a second magneticcircuit passing through the second section are equivalent. The first andsecond windings have winding directions opposite to each other.

More preferably, the power converter further includes a control deviceconfigured to control on/off of the plurality of switching elements soas to control an output voltage on a power line connected to the load.The control device controls a phase difference between a first carriersignal used for first pulse width modulation control for controllingpower conversion in a first power conversion path through which thefirst current flows and a second carrier signal used for second pulsewidth modulation control for controlling power conversion in a secondpower conversion path through which the second current flows such thatany critical point of one of the first and second currents coincideswith any critical point of the other one of the first and secondcurrents, and then generates signals for controlling on/off of theplurality of switching elements in accordance with the first and secondpulse width modulation control.

In still another aspect of the present invention, a power supply systemincludes a first DC power source, a second DC power source, and a powerconverter configured to execute DC power conversion between a power lineelectrically connected to a load and the first and second DC powersources. The power converter includes a plurality of switching elements,a first reactor, and a second reactor. The plurality of switchingelements are arranged to be included both in a first power conversionpath formed between the first DC power source and the power line and ina second power conversion path formed between the second DC power sourceand the power line. The first reactor is arranged to be included in thefirst power conversion path. The second reactor is arranged to beincluded in the second power conversion path. The first and secondreactors are formed integrally by a single magnetic component. Themagnetic component includes first and second windings electricallyconnected in series through which a first current flows, a third windingthrough which the second current flows, and a core. The core isconfigured to include a first section on which the first winding iswound, a second section on which the second winding is wound, and athird section on which the third winding is wound. The first to thirdwindings are wound on the first to third sections, respectively, suchthat, in a state where magnetic saturation does not occur in said core,(i) a magnetic flux produced from the first winding and flowing throughthe third winding and a magnetic flux produced from the second windingand flowing through the third winding counteract each other, when thefirst current flows through the first and second windings, and (ii)induced voltages produced in the first and second windings,respectively, by a magnetic flux produced from the third windingcounteract each other, when the second current flows through the thirdwinding.

Preferably, in the power supply system, the plurality of switchingelements include a first switching element electrically connectedbetween a first node and the power line, a second switching elementelectrically connected between a second node and the first node, a thirdswitching element electrically connected between a third nodeelectrically connected to a negative terminal of the second DC powersource and the second node, and a fourth switching element electricallyconnected between a negative terminal of the first DC power source andthe third node. The first reactor is electrically connected between apositive terminal of the first DC power source and the second node, andthe second reactor is electrically connected between the positiveterminal of the second DC power source and the first node.

More preferably, in the power supply system, the power converter isconfigured to be switchable, by controlling the plurality of switchingelements, between a first operation mode in which the DC powerconversion is executed with the first and second DC power sourceselectrically connected in series with the power line and a secondoperation mode in which the first and second DC power sources executethe DC power conversion to the power line in parallel.

Still more preferably, the power supply system further includes acontrol device configured to control on/off of the plurality ofswitching elements so as to control an output voltage on the power line.The control device controls a phase difference between a first carriersignal used for first pulse width modulation control for controllingpower conversion in the first power conversion path through which thefirst current flows and a second carrier signal used for second pulsewidth modulation control for controlling power conversion in the secondpower conversion path through which the second current flows such thatany critical point of one of the first and second currents coincideswith any critical point of the other one of the first and secondcurrents, and then generates signals for controlling on/off of theplurality of switching elements in accordance with the first and secondpulse width modulation control.

Further preferably, the phase difference is controlled such that one ofa rising edge and a falling edge of a first control pulse signalobtained by the first pulse width modulation control coincides with theother one of the rising edge and the falling edge of a second controlpulse signal obtained by the second pulse width modulation control.

Advantageous Effects of Invention

According to the present invention, two reactors respectively includedin current paths independently controlled in current are formedintegrally, thereby achieving reduction in size and weight of a magneticcomponent as well as a power converter and a power supply systemincluding the magnetic component.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a circuit diagram showing an exemplary configuration of apower converter and a power supply system including a magnetic componentin accordance with an embodiment of the present invention.

FIG. 2 is a typical control block diagram of the power supply systemshown in FIG. 1.

FIG. 3 is a diagram of operation waveforms of the power supply systemshown in FIG. 1.

FIG. 4 is a conceptual view illustrating a configuration of two reactorsformed as separate magnetic components as a comparative example.

FIG. 5 is a sectional view for illustrating a configuration of themagnetic component in accordance with an embodiment of the presentinvention.

FIG. 6 is a top view for illustrating the configuration of the magneticcomponent in accordance with an embodiment of the present invention.

FIG. 7 is an equivalent circuit diagram of the magnetic component inaccordance with an embodiment of the present invention.

FIG. 8 is a first conceptual view for illustrating magnetic behaviorwhen current flows through a first reactor.

FIG. 9 is a second conceptual view for illustrating magnetic behaviorwhen current flows through the first reactor.

FIG. 10 is a conceptual view for illustrating magnetic behavior whencurrent flows through a second reactor.

FIG. 11 is a block diagram showing a configuration of a power supplysystem 5 c in accordance with a second embodiment.

FIG. 12 includes circuit diagrams illustrating a first circuit operationin the parallel connection mode.

FIG. 13 includes circuit diagrams illustrating a second circuitoperation in the parallel connection mode.

FIG. 14 includes circuit diagrams illustrating DC/DC conversion (step-upoperation) for a first DC power source in the parallel connection mode.

FIG. 15 includes circuit diagrams illustrating DC/DC conversion (step-upoperation) for a second DC power source in the parallel connection mode.

FIG. 16 includes circuit diagrams illustrating a circuit operation inthe series connection mode.

FIG. 17 includes circuit diagrams illustrating DC/DC conversion (step-upoperation) in the series connection mode.

FIG. 18 is a block diagram showing an equivalent circuit viewed from theload side, in the parallel connection mode.

FIG. 19 is a diagram of waveforms representing an exemplary operationfor controlling the first power source.

FIG. 20 is a diagram of waveforms representing an exemplary operationfor controlling the second power source.

FIG. 21 is a table illustrating settings of control signals forswitching elements.

FIG. 22 is a diagram of waveforms representing PWM control in theparallel connection mode.

FIG. 23 is a diagram of conceptual waveforms representing relationshipbetween phase of reactor currents and magnetic flux densities.

FIG. 24 is a diagram of waveforms representing current phase achieved bycarrier phase control for reducing losses in the switching elements inthe parallel connection mode.

FIG. 25 includes circuit diagrams illustrating current paths in apredetermined period in FIG. 24.

FIG. 26 includes diagrams of current waveforms of the switching elementsin the current phase shown in FIG. 24.

FIG. 27 is a diagram of waveforms showing current phase when phasedifference between carrier signals is 0.

FIG. 28 includes diagrams of current waveforms of the switching elementsin the current phase shown in FIG. 27.

FIG. 29 is a table for illustrating carrier phase control in eachoperation state of the DC power sources.

DESCRIPTION OF EMBODIMENTS

In the following, embodiments of the present invention will be describedin detail with reference to the figures. In the figures, the same orcorresponding portions are denoted by the same reference characters and,basically, description thereof will not be repeated.

First Embodiment

FIG. 1 is a circuit diagram showing an exemplary configuration of apower converter and a power supply system including a magnetic componentin accordance with an embodiment of the present invention.

Referring to FIG. 1, power supply system 5 includes a DC power source10, a power converter 6, a DC power source 20, and a power converter 7.Power supply system 5 controls electric power supply from DC powersources 10, 20 to a load 30. Alternatively, power supply system 5charges DC power sources 10, 20 with electric power generated by load30.

In the present embodiment, DC power sources 10 and 20 are implemented bya power storage device such as a secondary battery or an electric doublelayer capacitor. DC power source 10 is implemented by, for example, asecondary battery such as a lithium ion secondary battery or anickel-metal hydride battery. DC power source 20 is implemented by, forexample, a DC voltage source element having superior outputcharacteristic such as an electric double-layer capacitor or a lithiumion capacitor. DC power sources 10 and 20 correspond to the “first DCpower source” and the “second DC power source”, respectively. However,it is also possible to implement DC power sources 10 and 20 by powerstorage devices of the same type.

Power converter 6 is connected between DC power source 10 and load 30.Power converter 7 is connected between DC power source 20 and load 30.It is understood that, in power supply system 5, DC power sources 10 and20 are connected in parallel to load 30 via power converters 6 and 7.

Load 30 operates receiving output voltage Vo from converters 6 and 7. Avoltage command value Vo* of output voltage Vo is set to a valueappropriate for the operation of load 30. The voltage command value maybe variably set in accordance with the state of load 30. Load 30 may beconfigured to be able to generate charging power for charging DC powersources 10 and 20 by regenerative power generation or the like. Forexample, load 30 is configured so as to include a traction motor for anelectric powered vehicle, such as a hybrid vehicle and an electricvehicle, and an inverter for controlling driving of that motor.

Power converter 6 executes bidirectional DC/DC conversion between DCpower source 10 and a power line PL connected to load 30. Powerconverter 7 executes bidirectional DC/DC conversion between DC powersource 20 and power line PL. Hereinafter, power converters 6 and 7 maybe referred to as converters 6 and 7, respectively.

Each of converters 6 and 7 has a configuration of a so-called step-upchopper circuit.

Specifically, converter 6 has power semiconductor switching elements(hereinafter briefly called “switching elements” as well) S1, S2 and areactor L1. Switching elements S1 and S2 are connected in series betweenpower line PL and a ground line GL. Reactor L1 is electrically connectedbetween the positive terminal of DC power source 10 and the connectionnode between switching elements S1 and S2.

Converter 7 includes switching elements S3, S4 and a reactor L2.Switching elements S3 and S4 are connected in series between power linePL and ground line GL. Reactor L2 is electrically connected between thepositive terminal of DC power source 20 and the connection node betweenswitching elements S3 and S4.

In the present embodiment, the switching elements can be implemented byan IGBT (Insulated Gate Bipolar Transistor), a power MOS (Metal OxideSemiconductor) transistor, a power bipolar transistor, or the like. Forswitching elements S1 to S4, anti-parallel diodes D1 to D4 are arranged.Switching elements S1 to S4 can be on/off controlled in response tocontrol signals SG1 to SG4 from a control device 40.

In converters 6 and 7, each implemented by a step-up chopper circuit, DCoutput is controlled in accordance with the duty ratio indicating theon-period ratio of an upper arm element (S1, S3) and a lower arm element(S2, S4) in a predetermined period (switching period). In general,switching elements S1 to S4 are controlled such that the upper armelement and the lower arm element are turned on/off in complementarymanner in accordance with a comparison between a DC signal indicatingthe duty ratio and a carrier signal of predetermined frequency.

The voltage conversion ratio (step-up ratio) in the step-up choppercircuit is known to be expressed by Equation (1) below, using voltage Viat the lower voltage side (DC power source side), output voltage VH atthe higher voltage side (load side) and duty ratio DT of the lower armelement. Here, duty ratio DT is defined by the on-period ratio of thelower arm element to the switching period which is the sum of the on-and off-periods of the lower arm element. The upper arm element isturned on in the off-period of the lower arm element.

VH=1/(1−DT)×Vi  (1)

Control device 40 is implemented by, for example, a CPU (CentralProcessing Unit) not shown and an electronic control unit (ECU) having amemory. Control device 40 is configured to perform arithmetic processingusing a detection value obtained by each sensor based on a map and aprogram stored in the memory. Alternatively, at least part of controldevice 40 may be configured to execute predetermined numeric and logicarithmetic processing by hardware, such as an electronic circuit.

Control device 40 generates control signals SG1 to SG4 that controlon/off of switching elements S1 to S4 in order to control output voltageVo to load 30. Although not shown in FIG. 1, detectors (voltage sensor,current sensor) are provided for detecting voltage (denoted by V[1]) andcurrent (denoted by I[1]) of DC power source 10, voltage (denoted byV[2]) and current (denoted by I[2]) of DC power source 20, and outputvoltage Vo. In converter 6, current I[1] is equivalent to current I(L1)flowing through reactor L1. Similarly, in converter 7, current I[2] isequivalent to current I(L2) flowing through reactor L2.

FIG. 2 is a typical control block diagram of power supply system 5 shownin FIG. 1.

Referring to FIG. 2, if control common to converters 6 and 7 (voltagecontrol of output voltage Vo) is executed simultaneously, there is apossibility of circuit failure. Therefore, converters 6 and 7 executeDC/DC conversion between DC power sources 10, 20 and load 30 such thatone of the batteries operates as a voltage source and the other batteryoperates as a current source.

Here, converter 6 shall control battery current I[1] in accordance witha current command value Ii* such that DC power source 10 operates as acurrent source. On the other hand, converter 7 controls output voltageVo in accordance with voltage command value Vo* such that DC powersource 20 operates as a voltage source.

Here, a relation represented by Equation (2) below holds among electricpower P[1] of DC power source 10, electric power P[2] of DC power source20, output power Po to load 30 and current command value Ii* of thecurrent source.

P[2]=Po−P[1]=Po−V[1]×Ii*  (2)

By setting current command value Ii* such that P[1]*=V[1]×Ii* is keptconstant in accordance with the detected value of voltage V[1] of DCpower source 10, electric power P[1] of DC power source 10 constitutingthe current source can be regulated to a power command value P[1]*.

It is also possible to exert control with DC power source 20 operatingas a current source and DC power source 10 operating as a voltagesource. In this case, by setting current command value Ii* such thatV[2]×Ii* is kept constant for electric power P[2] of DC power source 20constituting a current source, electric power P[2] of DC power source 20can be regulated in accordance with the power command value.

A current controller 41 controls the duty ratio of converter 6 such thatcurrent I[1] of DC power source 10 corresponds to current command valueIi*. Specifically, when a current deviation (Ii*−I[1]) is higher thanzero, the duty ratio is changed such that the on-period of the lower armelement (S2) becomes longer in order to increase current I[1]. Incontrast, when the current deviation (Ii*−I[1]) is lower than zero, theduty ratio is changed such that the on-period of the upper arm element(S1) of converter 6 becomes longer in order to reduce current I[1].

A voltage controller 42 controls the duty ratio of converter 7 such thatoutput voltage Vo corresponds to voltage command value Vo*. When avoltage deviation (Vo*−Vo) is higher than zero, voltage controller 42changes the duty ratio such that the on-period ratio of the lower armelement (S4) of converter 7 becomes larger in order to increase outputvoltage Vo. In contrast, when the voltage deviation (Vo*−Vo) is lowerthan zero, voltage controller 42 changes the duty ratio such that theon-period ratio of the upper arm element (S3) of converter 7 becomeslarger in order to reduce output voltage Vo.

In this way, DC power source 10 is subjected to current control byconverter 6 in accordance with current command value Ii*. On the otherhand, the output of DC power source 20 is controlled by converter 7 forvoltage control in accordance with voltage command value Vo*.

FIG. 3 shows exemplary operation waveforms of power supply system 5controlled by the control block shown in FIG. 2.

Referring to FIG. 3, operation waveforms in the case where P[1]>0 andP[2]>0 hold, DC power sources 10 and 20 discharge, and electric power issupplied to load 30 are shown. That is, output power Po expressed byPo=P[1]+P[2] is higher than zero.

Since current I[1] of DC power source 10 is controlled to be constant byconverter 6 in accordance with current command value Ii*, electric powerP[1] of DC power source 10 is also constant. Therefore, in the periodfrom time t1 to time t2 during which electric power of load 30 increaseswith voltage command value Vo* being constant, P[1] is maintainedconstant, while electric power P[2] of DC power source 20 increases.

In the period from time t2 to time t3, output power Po decreases, whilevoltage command value Vo* increases. Output voltage Vo is increased byconverter 7 in accordance with voltage command value Vo*. Furthermore,electric power P[1] is constant because current I[1] is controlled to beconstant by converter 6, while electric power P[2] gradually decreases.

In this way, electric power P[1] of DC power source 10 subjected tocurrent control is controlled in accordance with current command valueIi*. On the other hand, DC power source 20 will operate as a buffer forsupplying the difference between output power Po to load 30 and batteryelectric power P[1] while ensuring output voltage Vo.

As described above, in power supply system 5, current I(L1) flowingthrough reactor L1 of converter 6 and current I(L2) flowing throughreactor L2 of converter 7 are controlled independently. That is,converters 6 and 7 could no longer be controlled appropriately if aninduced voltage is produced in reactor L2 by current I(L1) or to thecontrary, an induced voltage is produced in reactor L1 by current I(L2).

If reactors L1 and L2 are formed integrally in accordance with PTL 1 orPTL 2, it will become difficult to independently control the respectivereactors in current due to interference of induced voltage. First, as acomparative example, to reliably avoid interference of induced voltage,a configuration in the case where reactors L1 and L2 are formed asindependent separate magnetic components will be described withreference to FIG. 4.

Referring to FIG. 4, a magnetic component 101 constituting reactor L1 isformed by a core 110 a and a winding 120 a wound on core 110 a. Gaps 112a are provided in core 110 a. Similarly, a magnetic component 102constituting reactor L2 is formed by a core 110 b and a winding 120 bwound on core 110 b. Gap 112 b are provided in core 110 b.

An inductance value L of the reactor is expressed by the number of turnsN of the coil, a magnetic resistance R of the magnetic material, and amagnetic resistance r of the gap, in accordance with Equation (3) below.

L=N×N/(R+r)  (3)

It is known that magnetic resistance R can be adjusted by the magneticproperty (relative permeability), size and shape (magnetic path lengthand cross sectional area) of cores 110 a and 110 b. Magnetic resistancer can be adjusted by the gap length and the number of gaps 112 a and 112b.

Since the magnetic material used for cores 110 a and 110 b has nonlinearcharacteristics, characteristics will be degraded due to a saturationphenomena if an excess magnetic flux is produced. Therefore, it isnecessary to design an effective sectional area S of the core such thata maximum magnetic flux density B(max) when a design maximum currentI(max) flows does not exceed the saturation magnetic flux density of thecore. B(max) is obtained by Equation (4) below.

B(max)=I(max)×N/(R+r)/S  (4)

In this way, when reactors L1 and L2 are formed by separate magneticcomponents 101 and 102, interference of induced voltage can be avoided,and a desired inductance value can be obtained while avoiding magneticsaturation because of the design of each magnetic component. That is,the inductance value can relatively easily be designed. On the otherhand, converters 6, 7 and power supply system 5 may be increased in sizedue to size increase of reactors L1 and L2 because two cores arerequired.

The present embodiment will describe a configuration for formingreactors L1 and L2 such that interference due to an induced voltage willnot occur and so as to be an integral magnetic component.

FIG. 5 is a schematic view illustrating a configuration of a magneticcomponent 100 in accordance with an embodiment of the present invention.FIG. 5 shows a cross sectional view of magnetic component 100.

Magnetic component 100 includes a core 150 and windings 121 a, 121 b and122. Windings 121 a and 121 b are electrically connected in series toconstitute the coil of reactor L1. Winding 122 constitutes the coil ofreactor L2.

Core 150 made of a magnetic material is formed to have leg portions 151,152 and 153. Gaps 161 to 163 are provided in leg portions 151 to 153,respectively. As described above, gaps 161 to 163 are useful foradjusting the inductance value.

FIG. 6 is a top view of magnetic component 100 shown in FIG. 5. Thearrangement relationship and the connection relationship among thewindings are shown in FIG. 6.

Referring to FIG. 6, winding 121 a is wound on leg portion 151 of core150, and winding 121 b is wound on leg portion 152 of core 150.

Windings 121 a and 121 b are electrically connected in series by aconducting wire 125 to form the coil of reactor L1. Winding 121 a and121 b connected in series are wound to have winding directions oppositeto each other. Winding 122 is wound on leg portion 153 of core 150 toform the coil of reactor L2.

In this way, in magnetic component 100, winding 121 a corresponds to the“first winding”, winding 121 b corresponds to the “second winding”, andwinding 122 corresponds to the “third winding.” Leg portion 151corresponds to the “first section”, leg portion 153 corresponds to the“second section”, and leg portion 152 corresponds to the “thirdsection.”

FIG. 7 is an equivalent circuit diagram of magnetic component 100.

Referring to FIG. 7, common current I(L1) flows through windings 121 aand 121 b of reactor L1. When current I(L2) flows through winding 122 ofreactor L2, an induced voltage Vm1a and an induced voltage Vm1b areproduced in windings 121 a and 121 b, respectively. Since windings 121 aand 121 b are wound in opposite winding directions, induced voltagesVm1a and Vm1b have opposite polarities and counteract each other. Aninduced voltage produced in winding 122 when current I(L1) flows throughreactor L1 is denoted by Vm2.

Magnetic behaviors when current I(L1) flows through reactor L1 will bedescribed with reference to FIGS. 8 and 9. FIG. 8 shows a magnetic fluxproduced by the current flowing through winding 121 a. FIG. 9 shows amagnetic flux produced by the current flowing through winding 121 b.

Referring to FIG. 8, when current I(L1) flows through winding 121 a, amagnetic flux 200 is produced by a magnetomotive force in accordancewith the product of current I(L1) and the number of turns of winding 121a.

Magnetic flux 200 is divided into a magnetic path 203 flowing throughleg portion 153 and a magnetic path 202 including leg portion 152. Inaccordance with the ratio of a magnetic resistance R2 of magnetic path202 to a magnetic resistance R3 of magnetic path 203, the ratio of themagnetic flux flowing through magnetic path 202 to the magnetic fluxflowing through magnetic path 203 is determined.

Referring to FIG. 9, current I(L1) flows through winding 121 b in commonwith winding 121 a. Accordingly, a magnetic flux 210 is produced by themagnetomotive force in accordance with the product of current I(L1) andthe number of turns of winding 121 b.

Magnetic flux 210 is divided into a magnetic path 211 flowing throughleg portion 151 and a magnetic path 212 flowing through leg portion 152.In accordance with the ratio of magnetic resistance R1 of magnetic path211 to magnetic resistance R2 of magnetic path 212, the ratio of themagnetic flux flowing through magnetic path 211 to the magnetic fluxflowing through magnetic path 212 is determined.

As is understood from FIGS. 8 and 9, in leg portion 152 where winding122 is wound, the magnetic fluxes flow through magnetic path 202 (FIG.8) and magnetic path 212 (FIG. 9) in the directions that counteract eachother. This is because windings 121 a and 121 b have opposite windingdirections.

Therefore, by designing such that the magnetic fluxes flowing throughmagnetic paths 202 and 212 are equal in strength, the magnetic fluxflowing through winding 122 produced by current I(L1) flowing throughreactor L1 can be made zero. Accordingly, induced voltage Vm2 producedin reactor L2 by current I(L1) flowing through reactor L1 can be madezero.

The magnetic flux produced by the current flowing through winding 122 isshown in FIG. 10.

Referring to FIG. 10, when current I(L2) flows through winding 122, amagnetic flux 220 is produced by a magnetomotive force in accordancewith the product of current I(L2) and the number of turns of winding122. Magnetic flux 220 is divided into a magnetic path 221 flowingthrough leg portion 151 and a magnetic path 223 including leg portion153. In accordance with the ratio of magnetic resistance R1 of magneticpath 221 and magnetic resistance R3 of magnetic path 223, the ratio ofthe magnetic flux flowing through magnetic path 221 and the magneticflux flowing through magnetic path 223 is determined.

Therefore, the magnetic fluxes flowing through magnetic paths 221 and223 become equal by forming leg portions 151 and 153 such that magneticresistances R1 and R3 are equivalent. At this time, induced voltagesVm1a and Vm1b having opposite polarities produced in windings 121 a and121 b, respectively, become equal in absolute value. As a result,induced voltage Vm1 of reactor L1 as a whole (Vm1=Vm1a+Vm1b) can be madezero.

In this way, in magnetic component 100, reactors L1 and L2 can be formedintegrally by windings 121 a, 121 b and 122 wound on common core 150.Furthermore, both induced voltage Vm2 produced in reactor L2 by currentI(L1) and induced voltage Vm1 produced in reactor L1 by current I(L2)can be made zero. That is, in magnetic component 100, two reactorsrespectively included in the current paths independently controlled incurrent can be formed integrally. Accordingly, reduction in size andweight of magnetic component 100 as well as converters 6 and 7 and powersupply system 5 including magnetic component 100 can be achieved.

The design of inductance value in magnetic component 100 will now bedescribed.

In magnetic component 100, as described with reference to FIG. 10, it isnecessary to design magnetic resistances R1 and R3 equally in order toequalize the magnitude of induced voltages Vm1a and Vm1b of windings 121a and 121 b produced by current I(L2).

Furthermore, the ratio of the inductance value of reactor L1 to theinductance value of reactor L2 can be designed in accordance with theratio of magnetic resistance R2 of magnetic paths 202, 212 including legportion 152 to magnetic resistances R1, R3. The ratio of the inductancevalue of reactor L1 to the inductance value of reactor L2 can bedesigned since the number of magnetic fluxes can also be adjusted inaccordance with the ratio of the number of turns N11 of winding 121 a(leg portion 151), the number of turns N13 of winding 121 b (leg portion153), and the number of turns N12 of winding 122 (leg portion 152). Inthis way, the inductance value of reactor L1 and the inductance value ofreactor L2 can be designed freely in accordance with the magneticresistances and the turns ratio. It is known that gaps 161 to 163 areuseful for adjusting the magnetic resistance of each magnetic path.

For example, if the ratio of magnetic resistances is designed asR1:R2=R3:R2=2:1 and the turns ratio is designed as N11=N12=N13, thecoupling factor between windings 121 a, 121 b and winding 122 will be0.33 (⅓), so that the inductance value of L1 can be designed to be twicethe inductance value of L2.

Moreover, for each of leg portions 151 to 153, the material of core 150and the shape and size of leg portions 151 to 153 can be designed suchthat magnetic saturation does not occur at the time when maximum currentI(max) in Equation (4) flows.

Second Embodiment

The second embodiment will describe another exemplary configuration of apower supply system to which magnetic component 100 described in thefirst embodiment is applied.

FIG. 11 is a block diagram showing a configuration of a power supplysystem 5 c in accordance with the second embodiment of the presentinvention.

Referring to FIG. 11, power supply system 5 c in accordance with thesecond embodiment includes DC power sources 10 and 20, a converter 50,and control device 40. As compared with power supply system 5 shown inFIG. 1, power supply system 5 c in accordance with the second embodimenthas a configuration provided with converter 50 instead of converters 6and 7. Converter 50 is connected between DC power sources 10, 20 andload 30. Converter 50 controls a DC voltage (output voltage Vo) on powerline PL connected to load 30 in accordance with a voltage command value.

Converter 50 includes switching elements S5 to S8 and reactors L3, L4.For switching elements S5 to S8, anti-parallel diodes D5 to D8 arearranged. Switching elements S5 to S8 are controlled on/off in responseto control signals SG5 to SG8 from control device 40.

Switching element S5 is electrically connected between power line PL anda node N1. Reactor L4 is connected between node N1 and a positiveterminal of DC power source 20. Switching element S6 is electricallyconnected between nodes N1 and N2. Reactor L3 is connected between nodeN2 and a positive terminal of DC power source 10. Switching element S7is electrically connected between nodes N2 and N3. Switching element S8is electrically connected between node N3 and ground line GL. Groundline GL is electrically connected to load 30 and a negative terminal ofDC power source 10.

As is understood from FIG. 11, converter 50 is configured to include astep-up chopper circuit for each of DC power sources 10 and 20.Specifically, for DC power source 10, a current bidirectional firststep-up chopper circuit, having an upper arm element formed by switchingelements S5, S6 and a lower arm element formed by switching elements S7,S8, is provided.

Similarly, for DC power source 20, a current bidirectional secondstep-up chopper circuit, having an upper arm element formed by switchingelements S5, S8 and a lower arm element formed by switching elements S6,S7, is provided. Switching elements S5 to S8 are included both in thepower conversion path between power source 10 and power line PL formedby the first step-up chopper circuit and the power conversion pathbetween DC power source 20 and power line PL formed by the secondstep-up chopper circuit.

As will be described in detail below, converter 50 is configured to beswitchable between a mode in which DC power sources 10 and 20 areconnected in parallel to load 30 to execute DC/DC conversion(hereinafter also referred to as a “parallel connection mode”), and amode in which DC power sources 10 and 20 are connected in series to load30 to execute DC/DC conversion (hereinafter also referred to as a“series connection mode”). In particular, converter 50 is capable ofoperating while switching between the parallel connection mode and theseries connection mode by controlling switching elements S5 to S8.

(Circuit Operation in Parallel Connection Mode)

The circuit operation in the parallel connection mode of converter 50will be described.

As shown in FIGS. 12 and 13, DC power sources 10 and 20 can be connectedin parallel with power line PL by turning on switching element S8 or S6.Here, in the parallel connection mode, equivalent circuit will differdepending on which is higher between voltage V[1] of DC power source 10and voltage V[2] of DC power source 20.

As shown at (a) of FIG. 12, when V[2]>V[1], by turning on switchingelement S8, DC power sources 10 and 20 are connected in parallel throughswitching elements S6 and S7. The equivalent circuit at this time is asshown at (b) of FIG. 12.

Referring to (b) of FIG. 12, between DC power source 10 and power linePL, by on/off control of switching element S7, the on-period and theoff-period of the lower arm element can be formed alternately.Similarly, between DC power source 20 and power line PL, by commonon/off control of switching elements S6 and S7, the on-period and theoff-period of the lower arm element of the step-up chopper circuit canbe formed alternately. Switching element S5 operates as a switch forcontrolling regeneration from load 30.

On the other hand, as shown at (a) of FIG. 13, when V[1]>V[2], byturning on switching element S6, DC power sources 10 and 20 areconnected in parallel through switching elements S7 and S8. Theequivalent circuit at this time is as shown at (b) of FIG. 13.

Referring to (b) of FIG. 13, between DC power source 20 and power linePL, by on/off control of switching element S7, the on-period and theoff-period of the lower arm element can be formed alternately.Similarly, between DC power source 10 and power line PL, by commonon/off control of switching elements S7 and S8, the on-period and theoff-period of the lower arm element of the step-up chopper circuit canbe formed alternately. Switching element S5 operates as a switch forcontrolling regeneration from load 30.

Next, referring to FIGS. 14 and 15, the voltage boosting (step-up)operation of converter 50 in the parallel connection mode will bedescribed.

FIG. 14 shows DC/DC conversion (step-up operation) for DC power source10 in the parallel connection mode.

Referring to (a) of FIG. 14, by turning on a pair of switching elementsS7 and S8 and by turning off a pair of switching elements S5 and S6, acurrent path 250 for storing energy in reactor L3 is formed. Thus, astate in which the lower arm element of the step-up chopper circuit ison is realized.

In contrast, referring to (b) of FIG. 14, by turning off the pair ofswitching elements S7 and S8 and by turning on the pair of switchingelements S5 and S6, a current path 251 for outputting the energy storedin reactor L3 with the energy of DC power source 10 is formed. Thus, astate in which the upper arm element of the step-up chopper circuit ison is realized.

By alternately repeating the first period in which the pair of switchingelements S7 and S8 is on and at least one of switching elements S5 andS6 is off and the second period in which the pair of switching elementsS5 and S6 is on and at least one of switching elements S7 and S8 is off,current path 250 of (a) of FIG. 14 and current path 251 of (b) of FIG.14 are formed alternately.

As a result, a step-up chopper circuit having the pair of switchingelements S5 and S6 as an equivalent of the upper arm element and thepair of switching elements S7 and S8 as an equivalent of the lower armelement is formed for DC power source 10. In the DC/DC convertingoperation shown in FIG. 14, there is no current circulation path to DCpower source 20 and, therefore, DC power sources 10 and 20 do notinterfere with each other. Specifically, power input/output to and fromDC power sources 10 and 20 can be controlled independently.

In such DC/DC conversion, the relation represented by Equation (5) belowholds between voltage V[1] of DC power source 10 and output voltage Voof power line PL. In Equation (5), Da represents the duty ratio of thefirst period in which the pair of switching elements S7 and S8 is on.

Vo=1/(1−Da)×V[1]  (5)

FIG. 15 shows DC/DC conversion (step-up operation) for DC power source20 in the parallel connection mode.

Referring to (a) of FIG. 15, by turning on a pair of switching elementsS6 and S7 and by turning off a pair of switching elements S5 and S8, acurrent path 260 for storing energy in reactor L4 is formed. Thus, astate in which the lower arm element of the step-up chopper circuit ison is realized.

In contrast, referring to (b) of FIG. 15, by turning off the pair ofswitching elements S6 and S7 and by turning on the pair of switchingelements S5 and S8, a current path 261 for outputting the energy storedin reactor L4 with the energy of DC power source 20 is formed. Thus, astate in which the upper arm element of the step-up chopper circuit ison is realized.

By alternately repeating the first period in which the pair of switchingelements S6 and S7 is on and at least one of switching elements S5 andS8 is off and the second period in which the pair of switching elementsS5 and S8 is on and at least one of switching elements S6 and S7 is off,current path 260 of (a) of FIG. 15 and current path 261 of (b) of FIG.15 are formed alternately.

As a result, a step-up chopper circuit having the pair of switchingelements S5 and S8 as an equivalent of the upper arm element and thepair of switching elements S6 and S7 as an equivalent of the lower armelement is formed for DC power source 20. In the DC/DC convertingoperation shown in FIG. 15, there is no current path including DC powersource 10 and, therefore, DC power sources 10 and 20 do not interferewith each other.

In such DC/DC conversion, the relation represented by Equation (6) belowholds between voltage V[2] of DC power source 20 and output voltage Voof power line PL. In Equation (6), Db represents the duty ratio of thefirst period in which the pair of switching elements S6 and S7 is on.

Vo=1/(1−Db)×V[2]  (6)

As described above, the current flowing through reactor L3 and thecurrent flowing through reactor L4 are controlled independently in theparallel connection mode of converter 50. As a result, powerinput/output to and from DC power sources 10 and 20 can be controlledindependently. That is, it is also necessary to form reactors L3 and L4such that an induced voltage is not produced in one reactor by thecurrent flowing through the other reactor.

(Circuit Operation in Series Connection Mode)

Next, referring to FIGS. 16 and 17, the circuit operation of converter50 in the series connection mode will be described.

As shown at (a) of FIG. 16, switching element S7 is fixed on, so that DCpower sources 10 and 20 can be connected in series to power line PL. Theequivalent circuit at this time is as shown at (b) of FIG. 16.

Referring to (b) of FIG. 16, in the series connection mode, between theseries-connected DC power sources 10 and 20 and power line PL, by commonon/off control of switching elements S6 and S8, the on-period and theoff-period of the lower arm element of the step-up chopper circuit canbe formed alternately. Switching element S5 is turned on in theoff-period of switching elements S6 and S8, thereby operating as aswitch for controlling regeneration from load 30. Further, by switchingelement S7 which is fixed on, a line 15 connecting reactor L3 toswitching element S8 is equivalently formed.

Next, referring to FIG. 17, the DC/DC conversion (step-up operation) inthe series connection mode will be described.

Referring to (a) of FIG. 17, switching element S7 is fixed on forconnecting DC power sources 10 and 20 in series, the pair of switchingelements S6 and S8 is turned on and switching element S5 is turned off.As a result, current paths 270 and 271 for storing energy in reactors L3and L4 are formed. As a result, for the series-connected DC powersources 10 and 20, a state in which the lower arm element of the step-upchopper circuit is on is realized.

Referring to (b) of FIG. 17, while switching element S7 is kept fixedon, the pair of switching elements S6 and S8 is turned off and switchingelement S5 is turned on, in contrast to (a) of FIG. 17. Thus, a currentpath 272 is formed. By current path 272, the sum of energy from DC powersources 10 and 20 connected in series and the energy stored in reactorsL3 and L4 is output to power line PL. As a result, for theseries-connected DC power sources 10 and 20, a state in which the upperarm element of the step-up chopper circuit is on is realized.

With switching element S7 kept fixed on, by alternately repeating thefirst period in which the pair of switching elements S6 and S8 is on andswitching element S5 is off and the second period in which switchingelement S5 is on and switching elements S6 and S8 are off, current paths270 and 271 of (a) of FIG. 17 and current path 272 of (b) of FIG. 17 areformed alternately.

In the DC/DC conversion in the series connection mode, the relationrepresented by Equation (7) below holds among voltage V[1] of DC powersource 10, voltage V[2] of DC power source 20 and output voltage Vo ofpower line PL. In Equation (7), Dc represents the duty ratio of thefirst period in which the pair of switching elements S6 and S8 is on.

Vo=1/(1−Dc)×(V[1]+V[2])  (7)

It is noted, however, that if V[1] and V[2] are different or if reactorsL3 and L4 have different inductances, reactors L3 and L4 come to havedifferent current values at the end of operation shown at (a) of FIG.17. Therefore, immediately after the transition to the operation shownat (b) of FIG. 17, if the current of reactor L3 is larger, a differencecurrent flows through a current path 273. If the current of reactor L4is larger, a difference current flows through a current path 274.

As described above, by controlling a plurality of switching elements S5to S8, converter 50 can use selectively the mode in which two DC powersources (batteries) 10 and 20 are connected in parallel and the mode inwhich the power sources are connected in series. As a result, it becomespossible to selectively use the parallel connection mode having improvedresponse to load power (supply of electric power to be consumed andreception of generated electric power) and improved manageability ofelectric power and the series connection mode having higher efficiencyand allowing higher usability of stored energy. Therefore, the two DCpower sources 10 and 20 can effectively be utilized.

In the parallel connection mode, reactors L3 and L4 as components ofconverter 50 need to be independently controlled in current. Therefore,by applying magnetic component 100 described in the first embodiment,reactors L3 and L4 can also be formed integrally. That is, for converter50 according to the second embodiment, reactors L3 and L4 are preferablyformed by applying magnetic component 100 according to the presentembodiment so as to achieve reduction in size and weight of the device.

Modification of Second Embodiment

As a modification of the second embodiment, a preferable controloperation in the parallel connection mode in the case where magneticcomponent 100 is applied to the power supply system described in thesecond embodiment will be described. The control operation describedbelow is achieved by hardware processing and/or software processing bycontrol device 40.

FIG. 18 shows an equivalent circuit viewed from the load side, in theparallel connection mode.

Referring to FIG. 18, in the parallel connection mode, a power sourcePS1 executing DC/DC conversion between DC power source 10 and load 30and a power source PS2 executing DC/DC conversion between DC powersource 20 and load 30 exchange power with load 30 in parallel. Powersource PS1 corresponds to the step-up chopper circuit executing theDC/DC converting operation shown in FIG. 14. Similarly, power source PS2corresponds to the step-up chopper circuit executing the DC/DCconverting operation shown in FIG. 15.

Power source PS1 has the DC/DC converting function with the voltageconversion ratio represented by Equation (5) between voltage V[1] of DCpower source 10 and output voltage Vo. Similarly, power source PS2 hasthe DC/DC converting function with the voltage conversion ratiorepresented by Equation (6) between voltage V[2] of DC power source 20and output voltage Vo.

In the parallel connection mode, similarly to converters 6 and 7 (FIG.1), if common control (voltage control of output voltage Vo) issimultaneously executed for both power sources, power sources PS1 andPS2 come to be connected in parallel on the side of the load and,therefore, there is a possibility of circuit failure. Therefore, one ofpower sources PS1 and PS2 operates as a voltage source controllingoutput voltage Vo. The other one of power sources PS1 and PS2 operatesas a current source regulating the current of the power source to acurrent command value.

Therefore, the voltage conversion ratio of each of power sources PS1 andPS2 is controlled such that the power source operates as a voltagesource or current source. For example, the power sources are controlledsuch that power source PS1 operates as a current source and power sourcePS2 operates as a voltage source like controlling the current in DCpower source 10 similarly to the first embodiment.

FIG. 19 is a diagram of waveforms representing a specific exemplaryoperation for controlling power source PS1 corresponding to DC powersource 10.

Referring to FIG. 19, a duty ratio Da (see Equation (5)) of power sourcePS1 is calculated in accordance with current feedback control for theoperation as the current source. In FIG. 19, a voltage signalrepresenting duty ratio Da is represented by the same referencecharacter Da.

A control pulse signal SDa of power source PS1 is generated by pulsewidth modulation (PWM) control based on a comparison between duty ratioDa and a periodical carrier signal 25 a. Generally, a triangular wave ora saw-tooth wave is used for carrier signal 25 a. The period of carriersignal 25 a corresponds to the switching frequency of each switchingelement, and the amplitude of carrier signal 25 a is set to a voltagethat corresponds to Da=1.0.

Control pulse signal SDa is set to a logic high level (hereinafterdenoted by H level) if the voltage indicating duty ratio Da is higherthan the voltage of carrier signal 25 a, and set to the logic low level(hereinafter denoted by L level) if the voltage is lower than thevoltage of carrier signal 25 a. The ratio of the H level period to theperiod of control pulse signal SDa (H level period+L level period), thatis, the duty ratio of control pulse signal SDa, is equivalent to Da.

A control pulse signal /SDa is an inversion signal of control pulsesignal SDa. When duty ratio Da becomes higher, the duty ratio of controlpulse signal SDa becomes higher. When duty ratio Da becomes lower, theduty ratio of control pulse signal SDa becomes lower.

Control pulse signal SDa corresponds to the signal for controllingon/off of the lower arm element of the step-up chopper circuit shown inFIG. 14. Specifically, the lower arm element is turned on in the H levelperiod, and the lower arm element is turned off in the L level period,of control pulse signal SDa. On the other hand, control pulse signal/SDa corresponds to the signal for controlling on/off of the upper armelement of the step-up chopper circuit shown in FIG. 14. The controloperation for power source PS1 shown in FIG. 19 corresponds to thecontrol operation by current controller 41 shown in FIG. 2.

FIG. 20 is a diagram of waveforms representing a specific exemplaryoperation for controlling power source PS2 corresponding to DC powersource 20.

Referring to FIG. 20, in power source PS2 also, by the PWM controlsimilar to that for power source PS1, a control pulse signal SDb and itsinversion signal /SDb are generated, based on a duty ratio Db (seeEquation (6)). The duty ratio of control pulse signal SDb is equivalentto Db, and the duty of control pulse signal /SDb is equivalent to(1.0−Db). Specifically, when duty ratio Db becomes higher, the H levelperiod of control pulse signal SDb becomes longer. On the contrary, whenduty ratio Db becomes lower, the L level period of control pulse signalSDb becomes longer.

Control pulse signal SDb corresponds to the signal for controllingon/off of the lower arm element of the step-up chopper circuit shown inFIG. 15. Control pulse signal /SDb corresponds to the signal forcontrolling on/off of the upper arm element of the step-up choppercircuit shown in FIG. 15.

Duty ratio Db is calculated in accordance with voltage feedback controlfor power source PS2 to operate as a voltage source. The controloperation for power source PS2 shown in FIG. 20 corresponds to thecontrol operation by voltage controller 42 of FIG. 2.

FIG. 21 shows settings of control signals for the respective switchingelements in the parallel connection mode.

Referring to FIG. 21, control signals SG5 to SG8 for controlling on/offof switching elements S5 to S8, respectively, are set based on thecontrol pulse signals (SGa, /SGa) for current control for power sourcePS 1 and the control signal pulses (SGb, /SGb) for voltage control forpower source PS2. Specifically, control signals SG5 to SG8 are set basedon a logical operation between control pulse signals (more specifically,in a mode of obtaining the logical sum).

Switching element S5 forms the upper arm element in each of the step-upchopper circuits shown in FIGS. 14 and 15. Therefore, control signal SG5controlling on/off of switching element S5 is generated by the logicalsum of control pulse signals /SDa and /SDb.

As a result, switching element S5 is on/off controlled such that itrealizes the functions of both the upper arm element of the step-upchopper circuit of FIG. 14 for controlling DC power source 10 and theupper arm element of the step-up chopper circuit of FIG. 15 forcontrolling DC power source 20.

Switching element S6 forms the upper arm element in the step-up choppercircuit of FIG. 14 and forms the lower arm element in the step-upchopper circuit of FIG. 15. Therefore, control signal SG6 controllingon/off of switching element S6 is generated in accordance with thelogical sum of control pulse signals /SDa and SDb. As a result,switching element S6 is on/off controlled such that it realizes thefunctions of both the upper arm element of the step-up chopper circuitof FIG. 14 and the lower arm element of the step-up chopper circuit ofFIG. 15.

Similarly, control signal SG7 for switching element S7 is generated inaccordance with the logical sum of control pulse signals SDa and SDb.Thus, switching element S7 is on/off controlled such that it realizesthe functions of both the lower arm element of the step-up choppercircuit of FIG. 14 and the lower arm element of the step-up choppercircuit of FIG. 15.

Further, control signal SG8 for switching element S8 is generated inaccordance with the logical sum of control pulse signals SDa and /SDb.Thus, switching element S8 is on/off controlled such that it realizesthe functions of both the lower arm element of the step-up choppercircuit of FIG. 14 and the upper arm element of the step-up choppercircuit of FIG. 15.

In the parallel connection mode, control signals SG6 and SG8 are set tocomplementary levels and, therefore, switching elements S6 and S8 areturned on/off in complementary manner. Accordingly, the operation whenV[2]>V[1] shown in FIG. 12 and the operation when V[1]>V[2] shown inFIG. 13 are switched naturally. Further, in each operation, switchingelements S5 and S7 are switched complementarily and, therefore, DC/DCconversion in accordance with duty ratios Da and Db can be executed inpower sources PS1 and PS2, respectively.

As described above, when operating converter 50 according to the secondembodiment in the parallel connection mode, PWM control is executed inparallel for each of DC power source 10 and DC power source 20. Thephase of carrier signals used for PWM control for DC power source 10 andDC power source 20 will now be described.

FIG. 22 shows a diagram of operation waveforms representing PWM controlin the parallel connection mode.

Referring to FIG. 22, carrier signal 25 a used for PWM control for DCpower source 10 and carrier signal 25 b used for PWM control for DCpower source 20 are periodical signals of the same frequency. Withcontrol according to the modification of the second embodiment, a phasedifference PH between carrier signals 25 a and 25 b is controlled. Inthe example of FIG. 22, phase difference PH=180°.

Control pulse signal SDa is generated based on a voltage comparisonbetween duty ratio Da calculated based on the voltage or current of DCpower source 10 and carrier signal 25 a. Similarly, control pulse signalSDb is obtained based on a comparison between duty ratio Db calculatedbased on the current or voltage of DC power source 20 and carrier signal25 b. Control pulse signals /SDa and /SDb are inversion signals ofcontrol pulse signals SDa and SDb.

Control signals SG5 to SG8 are set based on a logical operation ofcontrol pulse signals SDa (/SDa) and SDb (/SDb) in accordance with thelogic operation shown in FIG. 21. By turning on/off switching elementsS5 to S8 based on control signals SG5 to SG8, current I(L3) flowingthrough reactor L3 and current I(L4) flowing through reactor L4 arecontrolled as shown in FIG. 14. Current I(L3) corresponds to currentI[1] of DC power source 10, and current I(L4) corresponds to currentI[2] of DC power source 20.

From the principle of PWM control, even if phase difference PH isvaried, the length of the H level period of each of control pulsesignals SDa and SDb does not change. That is, currents I(L3) and I(L4)become equal in average value for the same duty ratios Da and Db,without depending upon phase difference PH. In this way, the outputs ofDC power sources 10 and 20 are controlled by duty ratios Da and Db, andthere is no influence exerted even if phase difference PH betweencarrier signals 25 a and 25 b is varied.

On the other hand, the phase relationship between control pulse signalsSDa and SDb changes in accordance with phase difference PH. Therefore,by changing phase difference PH between carrier signals 25 a and 25 b,the phase relationship (current phase) between current I(L3) and currentI(L4) changes.

FIG. 23 shows a diagram of conceptual waveforms representing therelationship between the phase of currents flowing through reactors andmagnetic flux densities.

Referring to FIG. 23, currents I(L3) and I(L4) of reactors L3 and L4change in phase in accordance with phase difference PH between thecarrier signals.

A magnetic flux density B(L3) of reactor L3 is proportional to currentI(L3), and a magnetic flux density B(L4) of reactor L4 is proportionalto current I(L4). A maximum magnetic flux density Bmax as the maximumvalue of a total magnetic flux density Bt (Bt=B(L3)+B(L4)) flowingthrough core 150 is proportional to the maximum value of currentI(L3)+I(L4).

Therefore, by controlling phase difference PH such that the relativemaximum point of current I(L3) and the relative minimum point of currentI(L4) agree in phase, the maximum value of current I(L3)+I(L4), that is,maximum magnetic flux density Bmax can be reduced. Alternatively,maximum value Bmax of total magnetic flux density can also be reduced bycontrolling phase difference PH such that the relative maximum point ofcurrent I(L4) and the relative minimum point of current I(L3) agree inphase.

In this way, when forming reactors L3 and L4 with magnetic component 100applied to converter 50, it is preferable from the viewpoint ofcontrolling the maximum value of magnetic flux density to control phasedifference PH such that the relative maximum point of the currentflowing through one reactor and the relative minimum point of thecurrent flowing through the other reactor agree in phase.

Also in power supply system 5 in accordance with the first embodiment,each of converters 6 and 7 can be controlled similarly to converter 50in the parallel connection mode. That is, similarly to the descriptionwith reference to FIGS. 18 to 20, 22, and 23, PWM control can beexecuted independently for each of converters 6 and 7, and the phasedifference between the carrier signals used in PWM control for the bothconverters can be controlled. Therefore, in magnetic component 100applied to power supply system 5 (FIG. 1), the maximum value of magneticflux density can be controlled by controlling phase difference PH suchthat the relative maximum point of one of currents I(L1) and I(L2) andthe relative minimum point of the other current agree in phase.

In converter 50 according to the second embodiment, when phasedifference PH between carrier signals is controlled so as to reduce themaximum value of magnetic flux density, it will also become possible toreduce losses in switching elements. Its description will be presentedbelow in detail.

First, as a typical example, control in the state where both of DC powersources 10 and 20 are in the power running state, that is, currentI(L3)>0 and current I(L4)>0 hold.

FIG. 24 is a diagram of waveforms illustrating current phase obtained byphase control according to the first embodiment for reducing losses inthe switching elements of converter 50 in the parallel connection mode.

Referring to FIG. 24, since switching elements S6 to S8 are on untiltime Ta, the lower arm element of the step-up chopper circuit is on foreach of DC power sources 10 and 20. Thus, both currents I(L3) and I(L4)increase.

At time Ta, switching element S6 is turned off, so that the lower armelement of the step-up chopper circuit is turned off for DC power source20. Thus, current I(L4) starts decreasing. Simultaneously with theturn-off of switching element S6, switching element S5 is turned on.

After time Ta, the lower arm element of the step-up chopper circuit isturned on for DC power source 10, and the lower arm element of thestep-up chopper circuit is turned off for DC power source 20. That is,current I(L3) increases, while current I(L4) decreases. At this time,the current path in converter 50 will be as shown at (a) of FIG. 25.

As is understood from (a) of FIG. 25, after time Ta, a differencecurrent between currents I(L3) and I(L4) will flow through switchingelement S8. That is, the current flowing through switching element S8decreases.

Referring to FIG. 24 again, when switching element S8 is turned off fromthe state after time Ta, the lower arm element of the step-up choppercircuit is turned off for DC power source 10. Thus, current I(L3) startsdecreasing. When switching element S6 is turned on, the lower armelement of the step-up chopper circuit is turned on for DC power source20. Thus, current I(L4) starts increasing again. That is, the currentpath in converter 50 changes from the state at (a) of FIG. 25 to thestate at (b) of FIG. 25. In the state at (b) of FIG. 25, the differencecurrent between currents I(L3) and I(L4) will flow through switchingelement S6, which means that the current flowing through switchingelement S6 decreases.

By turning off switching element S8 in the state at (a) of FIG. 25, thecurrent at the turn-off of switching element S8, that is, a switchingloss, can be reduced. By turning off switching element S6 in the stateat (b) of FIG. 25, the current at the turn-on of switching element S6,that is, a switching loss, can be reduced.

Here, the current phase, that is, phase difference PH between carriersignals 25 a and 25 b, is adjusted such that the decrease start timing(relative maximum point) of current I(L3) and the increase timing(relative minimum point) of current I(L4) agree in phase. Accordingly,at time Tb in FIG. 24, switching element S6 is turned on, and switchingelement S8 is turned off.

Referring to FIG. 24 again, at time Tc, switching element S5 is turnedoff, and switching element S8 is turned on. Accordingly, for each of DCpower sources 10 and 20, a state in which the lower arm element of thestep-up chopper circuit is on is realized. Accordingly, the state beforetime Ta described above is reproduced, and currents I(L3) and I(L4) bothincrease.

FIG. 26 shows the current waveforms of switching elements S6 and S8 inthe current phase shown in FIG. 24. The waveform of current I(S6) ofswitching element S6 is shown at (a) of FIG. 26, and the waveform ofcurrent I(S8) of switching element S8 is shown at (b) of FIG. 26.

Referring to (a) of FIG. 26, current I(S6) is represented as I(S6)=I(L4)in the period before time Ta and the period after time Tc. Sinceswitching element S6 is off in the period from time Ta to Tb, I(S6)=0holds. In the period from time Tb to Tc, I(S6)=−(I(L3)−I(L4)) holds asshown at (b) of FIG. 25.

Referring to (b) of FIG. 26, current I(S8) is represented as I(S8)=I(L3)in the period before time Ta and the period after time Tc. In the periodfrom time Ta to Tb, as shown at (a) of FIG. 25, I(S8)=−(I(L4)−I(L3))holds. Switching element S8 is off in the period from time Tb to Tc, andtherefore, I(S8)=0 holds.

FIG. 27 shows a current phase when phase difference PH between carriersignals is set at 0 with a duty ratio equivalent to FIG. 24 for acomparison with FIG. 24.

Referring to FIG. 27, when phase difference PH between carrier signals25 a and 25 b is 0, currents I(L3) and I(L4) will increase/decrease atdifferent timings (Tx, Ty, Tz, Tw), respectively.

Specifically, in the state before time Tx where switching element S5 isoff and switching elements S6 to S8 are on, current I(L3) and I(L4) bothincrease. By turning off switching element S8 at time Tx, current I(L3)starts decreasing. Simultaneously with the turn-off of switching elementS8, switching element S5 is turned on.

At time Ty, by turning off switching element S7, current I(L4) startsdecreasing. Simultaneously with the turn-off of switching element S7,switching element S8 is turned on. Accordingly, current I(L3) and I(L4)both decrease.

At time Tz, switching element S6 is turned off, and switching element S7is turned on. Accordingly, for DC power source 10, a state in which thelower arm element of the step-up chopper circuit is on is realized.Thus, current I(3) increases again. Furthermore, at time Tw, switchingelement S5 is turned off, and switching element S6 is turned on.Accordingly, the state before time Tx is reproduced, and current I(L3)and I(L4) both increase.

FIG. 28 shows the current waveforms of switching elements S6 and S8 inthe current phase shown in FIG. 27. The waveform of current I(S6) ofswitching element S6 is shown at (a) of FIG. 28, and the waveform ofcurrent I(S8) of switching element S8 is shown at (b) of FIG. 28.

Referring to (a) of FIG. 28, current I(S6) is represented as I(S6)=I(L4)in the period before time Tx and the period after time Tw. In the periodfrom time Tx to Ty, a current path similar to that of (b) of FIG. 25 isformed. Thus, I(S6)=−(I(L3)−I(L4)) holds. In the period from time Ty toTz, switching element S6 operates as the upper arm element for DC powersource 10. Thus, I(S6)=−I(L3) holds. In the period from time Ty to Tzduring which currents I(L3) and I(L4) both decrease, switching elementS6 operates as the upper arm element for DC power source 10. Thus,I(S6)=−I(L3) holds. In the period from time Tz to Tw, switching elementS6 is off. Thus, I(S6)=0 holds.

Referring to (b) of FIG. 28, current I(S8) is represented as I(S8)=I(L3)in the period before time Tx and the period after time Tw. In the periodfrom time Tx to Ty, switching element S8 is off. Thus, I(S8)=0 holds. Inthe period from time Ty to Tz during which currents I(L3) and I(L4) bothdecrease, switching element S8 operates as the upper arm element for DCpower source 20. Thus, I(S8)=−I(L4) holds. From time Tz to Tw, a currentpath similar to that of (a) of FIG. 25 is formed. Thus,I(S6)=−(I(L4)−I(L3)) holds.

From the comparison between current I(S6) generated at time Tb at (a) ofFIG. 26 and current I(S6) produced at time Tw at (a) of FIG. 28, it isunderstood that the turn-on current of switching element S6, that is,the switching loss at the turn-on, is reduced by adjusting phasedifference PH so as to achieve the current phase of FIG. 24.Furthermore, from the comparison between current I(S6) from time Tb toTc at (a) of FIG. 26 and current I(S6) from time Ty to Tz at (a) of FIG.28, it is understood that the conduction loss in switching element S6 isalso reduced.

Similarly, from the comparison between current I(S8) at time Tb at (b)of FIG. 26 and current I(S8) at time Tx at (b) of FIG. 28, it isunderstood that the turn-off current of switching element S8, that is,the switching loss at the turn-off, is reduced by adjusting phasedifference PH so as to achieve the current phase of FIG. 24.Furthermore, from the comparison between current I(S8) from time Ta toTb at (b) of FIG. 26 and current I(S8) from time Ty to Tz at (a) of FIG.28, it is understood that the conduction loss in switching element S8 isalso reduced.

In this way, by providing phase difference PH between carrier signals 25a and 25 b, losses in switching elements S5 to S8 can be reduced. Asshown in FIG. 24, in the state where DC power sources 10 and 20 are bothin the power running state, phase difference PH is set such that thedecrease start timing (relative maximum point) of current I(L3) and theincrease timing (relative minimum point) of current I(L4) agree inphase, that is, such that the turn-on timing of switching element S6coincides with the turn-off timing of switching element S8. Losses inswitching elements S5 to S8 can thereby be reduced. As a result, the DCpower conversion between DC power sources 10, 20 and power line PL (load30) can be executed efficiently. With such phase difference PH, the falltiming (or rise timing) of control pulse signal SDa and the rise timing(or fall timing) of control pulse signal SDb will coincide.

Control pulse signals SDa and SDb change in accordance with duty ratiosDa and Db. Therefore, it can be understood that phase difference PH thatcan achieve the current phase as shown in FIG. 24 also changes inaccordance with duty ratios Da and Db. Therefore, it is possible topreviously obtain the relationship between duty ratios Da, Db and phasedifference PH for reducing losses in the switching elements, and topreviously store that correspondence in control device 40 as a map(phase difference map) or a function expression (phase differencecalculation expression).

Then, in the PWM control for voltage/current control in DC power sources10, 20 in the parallel connection mode, phase difference PH for carrierphase control can be calculated based on calculated duty ratios Da andDb and in accordance with the phase difference map or the phasedifference calculation expression. Then, carrier signals 25 a and 25 bare produced such that they have calculated phase difference PH toexecute the PWM control. Accordingly, highly efficient DC powerconversion with reduced losses in switching elements S5 to S8 can beachieved.

While the state where both DC power sources 10 and 20 are in the powerrunning state has been described with reference to FIGS. 24 to 28,similar carrier phase control can also be executed in another state.

FIG. 29 is a table for illustrating carrier phase control in accordancewith the first embodiment of the present invention in each operationstate of the DC power sources.

Referring to FIG. 29, in a state A, both DC power sources 10 and 20 inthe power running state as described above. As shown in FIG. 24, phasedifference PH between the carrier signals is adjusted so as to achieve acurrent phase in which the decrease timing (relative maximum point) ofcurrent I(L3) and the increase timing (relative minimum point) ofcurrent I(L4) coincide at Tb in the drawing. Accordingly, the turn-onloss in switching element S6 and the turn-off loss in switching elementS8 at Tb can be reduced. Furthermore, as described above, the conductionloss in switching element S8 in the period from Ta to Tb and theconduction loss in switching element S6 in the period from Tb to Tc canbe reduced.

In a state B, both DC power sources 10 and 20 are in the regeneratingstate. In this state, phase difference PH between the carrier signals isadjusted so as to achieve a current phase in which the increase timing(relative minimum point) of current I(L3) and the decrease timing(relative maximum point) of current I(L4) coincide at Tb in the drawing.Accordingly, the turn-on loss in switching element S8 and the turn-offloss in switching element S6 at Tb can be reduced. Furthermore, asdescribed above, the conduction loss in switching element S6 in theperiod from Ta to Tb and the conduction loss in switching element S8 inthe period from Tb to Tc can be reduced.

In a state C, DC power source 10 is in the regenerating state, while DCpower source 20 is in the power running state. In this state, phasedifference PH between the carrier signals is adjusted so as to achieve acurrent phase in which the decrease timing of current I(L3) and thedecrease timing of current I(L4) coincide at Ta in the drawing.Accordingly, the turn-on loss in switching element S7 and the turn-offloss in switching element S5 at Ta can be reduced. Furthermore, asdescribed above, the conduction loss in switching element S5 in theperiod from Ta to Tb and the conduction loss in switching element S7 inthe period from Tc to Ta can be reduced.

In a state D, DC power source 10 is in the power running state, while DCpower source 20 is in the regenerating state. In this state, phasedifference PH between the carrier signals is adjusted so as to achieve acurrent phase in which the increase timing of current I(L3) and theincrease timing of current I(L4) coincide at Tc in the drawing.Accordingly, the turn-on loss in switching element S5 and the turn-offloss in switching element S7 at Tc can be reduced. Furthermore, asdescribed above, the conduction loss in switching element S5 in theperiod from Tb to Tc and the conduction loss in switching element S7 inthe period from Tc to Ta can be reduced.

In this way, in each of states A to D, by setting phase difference PHbetween the carrier signals such that any critical point (i.e., relativemaximum or minimum point) of one of currents I(L3) and I(L4) coincides(i.e., agrees in phase) with any critical point of the other current,losses in switching elements S5 to S8 can be reduced. Furthermore, it isunderstood that phase difference PH for reducing losses in switchingelements S5 to S8 differs depending on the combination of powerrunning/regenerating states of DC power sources 10 and 20. Therefore, inthe parallel connection mode of converter 50, it is preferable to setthe phase difference map or phase difference calculation expressiondescribed above for each combination of power running/regeneratingstates (states A to D in FIG. 29).

In this way, in control of converter 50 according to the modification ofthe present second embodiment, phase difference PH between carriersignals 25 a and 25 b is controlled in accordance with the operatingstate of converter 50, specifically, the duty ratio for current/voltagecontrol for DC power sources 10 and 20 or the duty ratio and the powerrunning/regenerating states of DC power sources 10 and 20.

In particular, in states A and B where both DC power sources 10 and 20are in the power running or regenerating state, phase difference PH iscontrolled such that the current phase shown in FIG. 29 is achieved inaccordance with the above-described phase difference map or phasedifference calculation expression, so that the relative maximum point ofone of currents I(L3) and I(L4) and the relative minimum point of theother current agree in phase. As described above, with such phasedifference PH, the fall timing (or rise timing) of control pulse signalSDa and the rise timing (or fall timing) of control pulse signal SDbwill coincide. Accordingly, in addition to reduction in losses dependingon switching, maximum magnetic flux density Bmax of magnetic component100 in which reactors L3 and L4 are formed integrally can be reduced.The cross sectional area of the core can thereby be made smaller, sothat further reduction in size and weight can be achieved.

As described above, the exemplary configurations of the converter (powerconverter) and the power supply system, including two reactors, to whichmagnetic component 100 according to the present embodiment is appliedhas been described by way of example in the first and secondembodiments. However, application of the present invention is notlimited to these power converter and power supply system. That is, themagnetic component according to the present embodiment is applicable toany circuit configuration that includes two reactors respectivelyincluded in current paths independently controlled in current.Accordingly, reduction in size and weight of the device can be achievedby forming the two reactors included in the power converter and thepower supply system integrally.

Further, it is noted that load 30 may be configured by any device thatoperates with controlled DC voltage Vo. Specifically, though examples inwhich load 30 is a traction motor or an inverter mounted on an electricvehicle or a hybrid vehicle have been described in the embodiments,application of the present invention is not limited to such examples.

It should be understood that the embodiments disclosed herein areillustrative and non-restrictive in every respect. The scope of thepresent invention is defined by the claims not by the description above,and is intended to include any modification within the meaning and scopeequivalent to the terms of the claims.

INDUSTRIAL APPLICABILITY

The present invention is applicable to a magnetic component with tworeactors included in different current paths as well as a powerconverter and a power supply system including the magnetic component.

1-11. (canceled)
 12. A power converter comprising: first and secondreactors configured by a magnetic component electrically connectedbetween a DC power source and a load; a plurality of switching elementsarranged between said DC power source and said load such that a firstcurrent flowing through said first reactor and a second current flowingthrough said second reactor are controlled independently; and a controldevice configured to control on/off of said plurality of switchingelements so as to control an output voltage on a power line connected tosaid load, said magnetic component including: first and second windingsconnected in series and forming said first reactor; and a third windingforming second reactor; and a core configured to include a first sectionon which said first winding is wound, a second section on which saidsecond winding is wound, and a third section on which said third windingis wound, said first to third windings being wound on said first tothird sections, respectively, such that, in a state where magneticsaturation does not occur in said core, a magnetic flux produced fromsaid first winding and flowing through said third winding and a magneticflux produced from said second winding and flowing through said thirdwinding counteract each other, when said first current flows throughsaid first and second windings, and induced voltages produced in saidfirst and second windings, respectively, by a magnetic flux producedfrom said third winding counteract each other, when said second currentflows through said third winding, and said control device controlling aphase difference between a first carrier signal used for first pulsewidth modulation control for controlling power conversion in a firstpower conversion path through which said first current flows and asecond carrier signal used for second pulse width modulation control forcontrolling power conversion in a second power conversion path throughwhich said second current flows such that any critical point of one ofsaid first and second currents coincides with any critical point of theother one of said first and second currents in phase, and then generatessignals for controlling on/off of said plurality of switching elementsin accordance with said first and second pulse width modulation control.13. The power converter according to claim 12, wherein in said core onwhich said first to third windings are wound, in a state where magneticsaturation does not occur in said core, a magnetic resistance of a firstmagnetic circuit passing through said first section and a magneticresistance of a second magnetic circuit passing through said secondsection are equivalent, when said second current flows through saidthird winding, and said first and second windings have windingdirections opposite to each other.
 14. The power converter according toclaim 12, wherein said control device controls said phase differencebetween said first carrier signal and said second carrier signal, whenboth said first and second currents flow in positive or negativedirection, such that the relative maximum point of one of said first andsecond currents and the relative minimum point of the other of saidfirst and second currents agree in phase.
 15. A power supply systemcomprising: a first DC power source; a second DC power source; and thepower converter as defined in claim 12, said power converter beingconfigured to execute DC power conversion between a power lineelectrically connected to a load and said first and second DC powersources, said plurality of switching elements including: a firstswitching element electrically connected between a first node and saidpower line; a second switching element electrically connected between asecond node and said first node; a third switching element electricallyconnected between a third node electrically connected to a negativeterminal of said second DC power source and said second node; and afourth switching element electrically connected between a negativeterminal of said first DC power source and said third node, said firstreactor being electrically connected between a positive terminal of saidfirst DC power source and said second node, and said second reactorbeing electrically connected between the positive terminal of saidsecond DC power source and said first node.
 16. The power supply systemaccording to claim 15, wherein in said core on which said first to thirdwindings are wound, in a state where magnetic saturation does not occurin said core, a magnetic resistance of a first magnetic circuit passingthrough said first section and a magnetic resistance of a secondmagnetic circuit passing through said second section are equivalent,when said second current flows through said third winding, and saidfirst and second windings have winding directions opposite to eachother.
 17. The power supply system according to claim 15, wherein saidpower converter is configured to be switchable, by controlling saidplurality of switching elements, between a first operation mode in whichsaid DC power conversion is executed with said first and second DC powersources electrically connected in series with said power line and asecond operation mode in which said first and second DC power sourcesexecute said DC power conversion to said power line in parallel.
 18. Thepower supply system according to claim 15, wherein said control devicecontrols said phase difference between said first carrier signal andsaid second carrier signal, when both said first and second DC powersources are in the power running or regenerating state, such that therelative maximum point of one of said first and second currents and therelative minimum point of the other of said first and second currentsagree in phases.
 19. The power supply system according to claim 15,wherein said phase difference is controlled such that one of a risingedge and a falling edge of a first control pulse signal obtained by saidfirst pulse width modulation control coincides with the other one of therising edge and the falling edge of a second control pulse signalobtained by said second pulse width modulation control.